1. Field of the Invention
This invention relates generally to a wideband power amplifier and, more particularly, to a capacitively coupled, wideband MMIC power amplifier that employs an active feedback regulator to provide reduced DC and RF process variation dependence to make the amplifier more easily and readily manufacturable.
2. Discussion of the Related Art
Wideband MMIC power amplifiers are widely employed in the telecommunications industry. An MMIC amplifier which can deliver high power and produce large voltage swings over a wide frequency bandwidth is important for many telecommunication applications, such as high data rate fiber transmission systems, microwave frequency converter applications, etc. For example, there exists a need in the art for a low-cost, wide bandwidth MMIC amplifier that can deliver medium to high power ( greater than 12 dBm) at an output voltage greater than 2 volts peak-to-peak over several decades of frequency bandwidth (20 kHz-20 GHz). However, making a semiconductor amplifier that is both wideband and high power is difficult because of certain manufacturing limitations, particularly the limitations of DC and RF process variation dependence.
Various circuit designs have been proposed in the art to make wide bandwidth, high power semiconductor amplifiers. For example, a capacitively coupled distributed amplifier has been proposed that is one of the best known designs for achieving this goal. In one particular known MESFET MMIC design, a series capacitance is employed in connection with the gate of each FET to reduce the effective shunt capacitance used to synthesize the input transmission line of the distributed amplifier to enable a greater distributed amplifier bandwidth without scaling down the size of the power FETs used. It has been shown that this approach could achieve 1 W of output power over a 2-8 GHz bandwidth. This circuit design has been extended to a capacitively coupled HBT distributed amplifier to improve the power added efficiency (PAE) and linearity of the wideband power amplifier. In this modified design, 0.5 W was obtained over a 2-8 GHz bandwidth. In both of the designs, the lower frequency band edge is limited to 2 GHz due to the capacitive coupling technique, thus, preventing them from being used in some telecommunications applications requiring a lower frequency band.
FIG. 1 is a schematic diagram of a conventional capacitively coupled power distributed amplifier 10 that is known in the art. The amplifier 10 includes an input transmission line 12 and an output transmission line 14, where input inductors 16 are periodically connected in series along the input transmission line 12 and output inductors 18 are periodically connected in series along the output transmission line 14. The input line 12 and the output line 14 can be traces on a printed circuit board. An input signal, such as a microwave signal, applied to an input node 20 of the input transmission line 12 is electrically coupled into the output transmission line 14, and is provided at an output node 22 of the output transmission line 14. An input termination resistor 26 is provided at an opposite end of the transmission line 12 from the node 20 to prevent back reflections on the transmission line 12 that may act to reduce the input signal depending on the relative phase of the reflection. Likewise, an output termination resistor 28 is provided at an end of the output transmission line 14 opposite to the node 22 to prevent back reflections of the output signal on the transmission line 14.
The amplifier 10 includes a plurality of amplifier stages 32 that are distributed between the inductors 16 and 18 along the transmission lines 12 and 14, and act to couple electromagnetic energy from the input transmission line 12 to the output transmission line 14 with a certain amount of gain. Each amplifier stage 32 includes an amplifying device 34 that may be an HBT transistor. The amplifier 10 defines a distributed transmission line modeled by the series inductors 16 and 18 and a shunt capacitance Cxcfx80 in the amplifying devices 34. The bandwidth of the signal being coupled from the input transmission line to the output transmission line 14 is determined by the inductance of the inductors 16 and 18 and the shunt capacitance Cxcfx80. Although the inductors 18 and the output shunt capacitance of the amplifying devices 34 affects the output power on the output transmission line 14, it is typically the input shunt capacitance Cxcfx80 of the amplifying devices 34 that affects the overall gain-bandwidth product. Therefore, the practical upper frequency bandwidth limit of the distributed amplifier 10 is usually determined by the cut-off frequency fci of the input distributed transmission line 12.
The cut-off frequency fci is defined as:
fci=1/(xcfx80Lxcex93Cxcex93)xe2x80x83xe2x80x83(1)
where Lxcex93 is the inductance of the inductors 16 and Cxcex93 is the effective shunt capacitance of the amplifier stages 32. To increase the gain and output power of the amplifier 10, it is necessary to increase the size of the amplifying devices 34, or increase the bias current applied to the transistor in the devices 34. However, when the amplifying devices 34 are biased with more current, a higher input diffusion capacitance is created in the amplifying devices 34. This diffusion capacitance causes the amplifying devices 34 to appear to have a large shunt capacitance Cxcfx80, which acts to reduce the cut-off frequency as defined in equation (1). So, as the power output of the amplifier 10 increases, the bandwidth typically decreases.
To overcome this upper bandwidth limitation, it is known in the art to employ a series capacitor 36 in combination with each amplifying device 34 in each stage 32. The series capacitor 36 acts as a division of the shunt capacitance Cxcfx80 in the amplifying devices 34 that reduces the input capacitance. Because the capacitor 36 is in series with the shunt capacitance Cxcfx80 in the amplifying devices 34, the effective capacitance of the transconductance of the amplifier 10 can be reduced, thus increasing the upper bandwidth limitation for high transition bias currents. This allows a designer to develop a distributed amplifier with a greater upper bandwidth cut-off frequency. In this design, the effective shunt capacitance Cxcex93 is Cbb Cxcfx80/(Cbb+Cxcfx80), where Cbb is the capacitance of the capacitor 36. It is desirable to have a low value for Cbb to produce a small Cxcex93 which allows a greater bandwidth without changing the output characteristics of the amplifier 10. Thus, a wider bandwidth can be achieved without sacrificing power.
Because the capacitor 36 has an infinite or high impedance at DC or low frequencies, the low end of the frequency bandwidth is limited. To overcome this limitation, it is known to include a shunt resistor 38 in parallel with the capacitor 36 to provide a signal path around the capacitor 36 for low frequency or DC signals. The low frequency performance of the amplifier 10 is determined by:
f=xc2xdxcfx80RbbCbbxe2x80x83xe2x80x83(2)
where Rbb is the value of the resistor 38. It is thus desirable to provide a high resistance for the resistor 38 to get a low frequency response at the lower end of the bandwidth. The lower frequency band edge is approximately determined by the pole produced by the capacitor 36 and the resistor 38. The resistor 38 allows a bias to the base or gate terminal of the amplifying devices 34. As will be shown below, a large value for the resistor 38 causes the manufactured amplifier to be more sensitive for variations in process.
A benefit of the capacitive coupling technique as discussed above is that the upper frequency bandwidth can be extended for a given output power, or, for a greater bandwidth, the device periphery can be increased to be obtain higher output power. The net gain is an increase in power bandwidth. However, this increase in power bandwidth is at the expense of gain because of the capacitive voltage division at the input of the amplifier 10, and the lower frequency response. Because the capacitor 36 is usually chosen to be quite small in order to raise the upper frequency band edge, the resistor 38 must be large ( greater than 1 kxcexa9) in order to extend the lower frequency response. For a typical value of the resistor 38 of 400xcexa9 the lower frequency band edge is around 2 GHz for a practical HBT based design. This is unacceptable for use in wideband fiber telecommunications which requires a low end frequency response to be around 20 kHz and advanced broadband RF switch applications with similar low frequency requirements.
FIG. 2 shows the effect of the resistor 38 on the low frequency response of a GaAs HBT capacitively coupled distributed amplifier, such as the amplifier 10. This graph shows that a large value for the resistor 38 of 10 kxcexa9 is required to achieve a flat gain response down to 20 kHz. Thus, a large value for the resistor 38 in the Kilo-ohms range is required in order to extend the practical low frequency response for such applications. However, due to the finite base currents of the HBT devices, which can vary by a factor of 5-10 due to the HBT DC beta process variation in a high volume production line, a large value for the resistor 38 can induce large bias variations in the amplifier 10 which consequently results in a large unacceptable RF performance variation of the MMIC product. DC beta is an HBT parameter that is the forward current gain of the HBT and is approximately equal to the collector current divided by the base current of the HBT.
FIG. 3 is a graph with power on the horizontal axis and current on the vertical axis that shows the amplifier bias current sensitivity to the HBT DC beta process variations. This graph shows that 70% variations in total amplifier bias current Ice can result from a typical HBT DC beta variation of 200-1000. The commercial wafer acceptance criteria for DC beta is usually set very low (150), and this is only a minimum DC beta requirement with an unbounded upper range. Because of the complex process dependence, no upper bound restrictions on the DC beta can be imposed. Therefore, it is necessary that the resistor 38 have a high value for the reasons discussed above, but this high value causes problems in the manufacturability of the resistor by making it more sensitive to variations in the manufacturing process.
The resulting impact on amplifier output performance is shown in FIG. 4 which illustrates the output power, gain and power added efficiency (PAE) of the amplifier 10, vs. input power dependence on the DC beta. The large bias sensitivity illustrated in FIG. 3 has resulted in large RF sensitivity of between 2-2.5 dB for gain and output power. This type of variation is unacceptable in commercial telecommunications where large emphasis exists on plug and replace sockets for low cost field maintenance.
What is needed is a technique for increasing the bandwidth and power output for an HBT power distributed amplifier, that has application for fiber optic telecommunications, without sacrificing existing process variations. It is therefore an object of the present invention to provide such an amplifier.
In accordance with the teachings of the present invention, a capacitively coupled power distributed amplifier is disclosed that provides a high power output and a wide bandwidth necessary for telecommunications applications without being sensitive to process variations. The amplifier includes an input load termination resistor opposite an input end of an input transmission line, and an output load termination resistor opposite an output end of a output transmission line. A series of distributed amplifying devices are connected between the input transmission line and the output transmission line.
An active feedback regulation loop is connected between the output load termination resistor and the input load termination resistor. The feedback loop includes an active feedback regulator connected across the output load resistor, and a lossy, low-pass distributed transmission line filter connected between the regulator and the input load termination resistor. The regulator senses a voltage potential on the output load termination resistor, compares the voltage potential to a reference potential, and provides a regulated output signal to bias the amplifying devices to control their output in a desirable manner. The transmission line filter filters out noise for the long connection between the input load resistor and the output load resistor. The regulator and the distributed transmission line filter combine to provide a reduction in DC bias current sensitivity, RF gain and output power sensitivity over wide process variations for a wide range of DC betas.
The invention as described herein provides a number of novel features over the known capacitively coupled distributed amplifier. These features include using the output load termination resistor as a DC sensor resistor for the regulator; directly connecting the regulator to an RF sensitive node of the amplifiers output transmission line; using an RF blocking resistor for inhibiting the broadband dynamic loading effects of the regulator from the RF sensitive node; using a compensation capacitor on a high impedance node of the regulator to allow a reasonable size monolithic capacitor for filtering out the RF signal introduced into the regulator from the output of the amplifier; and using the lossy, distributed low-pass filter transmission line for creating a controlled broadband feed from the output of the regulator to the input load termination resistor to inhibit in-band resonances caused by the interaction with an off-chip by-pass load network.
Additional objects, features and advantages of the present invention will become apparent from the following description and appended claims, taken in conjunction with the accompanying drawings.